In a spread spectrum communication system, wherein transmitted signal power is spread over a transmission bandwidth many times greater than the information bandwidth, signal power is sacrificed in an effort to achieve jam resistance, reduce the probability of detection, and to provide multipath signal rejection. Typically, spread spectrum communication systems employ a pair of pseudo-noise (PN) sequence generators at the transmitting and receiving stations, to respectively control the modulation and demodulation of the phase of the carrier signal. At the transmitter, a PN bit sequence may be encoded to produce a wide band signal which is combined with a relatively narrow bandwidth digital data signal so as to form modulation signals. The modulation signals may be applied to a multiphase (e.g. quad phase) modulator through which the carrier signal is phase shift keyed (PSK) modulated, to thereby obtain a spread spectrum signal having a wide bandwidth over which the power density is thinly spread. FIG. 1a shows an exemplary spectrum characteristic of a PSK signal, the transmitted signal power of which is confined to a prescribed frequency, while FIG. 1b depicts an exemplary spectrum characteristic of a PSK signal, the transmitted signal power of which is spread over a fairly large bandwidth as a result of subjecting the original PSK signal to spread spectrum transmission. FIG. 1c illustrates the spectrum characteristic of a spread spectrum transmitted PSK signal as shown in FIG. 1b in the presence of a jamming signal occupying some fixed or narrow bandwidth region of the spectrum. As will be readily appreciated from a comparison of FIGS. 1a through 1c, whereas the presence of a jamming signal having a frequency which corresponds to that of a transmitted non-spread PSK signal would effectively prevent intelligent detection of the non-spread PSK signal, such a jamming signal does not prevent acquisition of a spread spectrum PSK signal because the information contained in the signal is spread out over a large number of frequencies exclusive of that of the jamming signal.
To recover the original narrow bandwidth data signal from a transmitted spread spectrum signal, the receiver correlates a PN sequence or code with the incoming spread spectrum signal. This PN code generated at the output of the PN sequence generator at the receiver matches or is identical to the PN sequence employed at the transmitter for the modulation of the phase of the carrier signal, so that when the PN sequence generated at the receiver is synchronized with the PN sequence contained in the received spread spectrum signal, correlation of the two signals will enable demodulation and data recovery to be achieved.
For the purpose of synchronizing the PN sequence generated at the receiver with that contained in the incoming spread spectrum signal, timing control circuitry is provided in the receiver which controls the clock rate and generation of the receiver's PN code in an effort to maintain full correlation between the two PN sequences. The degree of synchronization must be such that the local PN code falls within .+-.1 baud period of the PN code contained in the received spread spectrum signal.
FIG. 2 illustrates the power distribution characteristics for the correlation of a pair of PN codes so employed in a spread spectrum communication system. As can be seen from FIG. 2, the power contained within the correlation of the PN codes is at a maximum when the codes are exactly in phase and decreases substantially as the phase difference or time differential between the PN code generated at the receiver and that contained in the receiver spread spectrum signal shifts towards .+-.1 baud period. Outside the .+-.1 baud period time differential there is effectively no useful correlation between the two PN codes, so that it is of paramount inportance that precise synchronization between the two codes be maintained in order that data recovery can be achieved.
One approach to synchronizing the two codes, referred to as the "dither" technique, involves a scheme whereby the clock phase by which the locally generated PN sequence is controlled is scanned (retarded or advanced) and the amount of correlation for the two phases (advanced phase and retarded phase) is compared. Namely, the locally generated PN code is "dithered" in time and the difference between the results of the correlations for a pair of advanced and retarded PN codes with the received PN code is used to direct a local oscillator towards a frequency which will correct for phase error between the locally generated and received PN codes. A general description and illustration of a "dithering" technique may be found in U.S. Pat. No. to Gordy et al 4,017,798 and reference may be had thereto for further understanding of such a technique as the case requires. Now, although a "dithering" scheme, as described briefly above and represented, for example, by the system described in the above-cited Gordy et al patent, is fairly simple, since only a single channel is required, it is still subject to jamming. For example, since alternating signal measurements are carried out for the advanced and retarded locally generated PN code, it is possible to introduce noise into, or spoof, the system so as to effectively offset one or more measured values in a sequence and thereby cause an improper shift of the local oscillator. Also, a dithering scheme suffers inherently higher tracking jitter due to noise than does a delay lock loop.
To overcome the problem of spoofing and degraded noise performance, to which a "dithering" scheme is vulnerable, there has been developed a delay lock loop code tracking system, an exemplary configuration of which is illustrated in FIG. 3. In accordance with a delay lock loop scheme, signal measurements are carried out for a pair of channels simultaneously, the phases of the two channels being relatively offset and the locally generated PN code is controlled so as to maintain the relative phase difference between the two channels at a constant value. In the system shown in FIG. 3, an incoming spread spectrum IF signal containing the transmitted PN code with which the locally generated PN code is to be synchronized is distributed from an input terminal 11 to a pair of mixers 12 and 13. In each of mixers 12 and 13, the received signal is combined with a PN signal code locally produced by PN code generator 17, local oscillator 16, and quadphase modulator 15. To provide a relative phase offset between the codes applied to mixers 12 and 13, a delay 14 is inserted in the signal path between the output of quad-phase modulator 15 and mixer 13. The outputs of mixers 12 and 13 are filtered in IF filters 18 and 19 and applied via a pair of variable gain amplifiers 20 and 21 to envelope detectors 24 and 25 respectively. The outputs of envelope detectors 24 and 25 represent the degree of correlation of the PN code contained in the received spread spectrum signal with the pair of relatively offset PN codes generated at the receiver for channels A and B.
FIG. 4 shows a pair of power distribution characteristics, like the one shown in FIG. 2, corresponding to the signal power correlations for channels A and B, respectively, when the system shown in FIG. 3 is properly controlling the timing of the locally generated PN codes and the gains of the channels A and B are equal. More specifically, the clock control input of PN generator 17 is supplied from voltage controlled clock 22, the input of which is coupled to the output of a tracking loop filter 26. The input of tracking loop filter 26 is coupled to the difference output port of a sum and difference circuit 27. The sum output port of sum and difference circuit 27 is coupled through an AGC loop filter to the gain control inputs of variable gain amplifiers 20 and 21. Now, the output of voltage controlled clock 22 is set such that the PN sequence produced at the output of quad-phase modulator 15 leads or is advanced relative to the PN code contained in the signal at terminal 11 by a prescribed time differential +.DELTA.T/2. The delay imparted by delay 14 is equal to a time differential .DELTA.T so that the PN code appearing at the output of delay 14 lags or is retarded relative to the PN code contained in the signal at terminal 11 by the time differential -.DELTA.T/2. Therefore, for equal gains through the two channels A and B, the outputs of envelope detectors 24 and 25 will be equal when the locally generated PN code inputs to mixers 12 and 13 are equally shifted about the maximum correlation timing position (0) shown in FIG. 4a. Accordingly, with this arrangement, either channel A or channel B may be used for data recovery provided that the correlation loss resulting from the phase shift is tolerable, or a third channel may be coupled to the output of quad-phase modulator 15 through a delay having phase offset of .DELTA.T/2 to provide the proper code phasing for full signal correlation.
Now, compared to the "dithering" technique described previously, for a delay between the two channels equal to the peak-to-peak time dither, the delay lock loop has inherently less tracking jitter than the dithered loop. However, the principal difficulty with the delay lock loop scheme is the need to accurately match the IF gains in the two channels A and B, as a one dB gain mismatch may cause a one-quarter loss in correlation. Namely, although AGC loop filter 23 is employed to control the gain of variable gain amplifier 20 and 21 for the respective channels, the characteristics of the separate channel components may result in a gain mismatch so that, even if the locally generated PN codes are properly timed relative to the PN code contained in the received spread spectrum signal, the outputs of envelope detector 24 and 25 may not be equal, causing a non-zero difference signal to be produced by sum and difference circuit 27, the sign or polarity of which is dependent upon the gains of the two channels A and B, thereby causing an erroneous shift of the output of clock 22 and a resultant mistiming of the PN codes.